Input capacitance
Phil Hobbs
more, perhaps somebody here can enlighten me.
Back in the old days, there was something called an 'input capacitance'.
You changed the voltage on an input, it cost you a bit of charge. No
worries.
Nowadays there are 'differential' and 'common mode' input capacitances.
Still no huge problem--an unbalanced input signal is half differential
and half common-mode, so effectively Cin = 0.5[Cin(diff) + Cin(CM)].
But some parts have hugely different CM and differential capacitance,
e.g. the OPA657, whose Cin is 0.7 pF differential but 4.5 pF CM!
Anybody know the origin of the big difference?
Cheers,
Phil Hobbs
John Larkin
Probably the ESD diodes add the cm capacitance, and they're measuring
the differential capacitance 3-terminal style.
2.25
-------+--------||-----------
| |
_|_ |
___ 0.7 |
| |
-------+--------||-----------
2.25 |
|
|
rails
John
cassiope
I don't really "know", but my speculation is: the CM capacitance is
mostly
stray C. After all, neither is very large. The DM capacitance is
small because
the actual JFETs are tiny, and probably cascoded.
Yes, it's a real shame that even the simplified schematics no longer
are
part of most datasheets :(
-f
Jim Thompson
Specsmanship ;-)
Ignoring the well-to-substrate capacitance allows one to "spec" the
tiny differential part.
It's always been there ;-)
...Jim Thompson
--
| James E.Thompson, P.E. | mens |
| Analog Innovations, Inc. | et |
| Analog/Mixed-Signal ASIC's and Discrete Systems | manus |
| Phoenix, Arizona 85048 Skype: Contacts Only | |
| Voice:(480)460-2350 Fax: Available upon request | Brass Rat |
| E-mail Icon at http://www.analog-innovations.com | 1962 |
I love to cook with wine Sometimes I even put it in the food
mi...@sushi.com
It could be a function of input bias cancellation circuitry.
Note that input capacitances are usually GBD (guaranteed by design)
parameters. That is, assuming wafer parametric test is satisfactory,
the capacitance limit should be valid. Expect the value to be guard-
banded heavily unless it is very important to the application.
Phil Hobbs
Hmm, that's probably part of it, true.
I think there's more circuity stuff down underneath there though...the
OPA656, which is the unity-gain stable version, specs 2.8 pF CM.
I wish I knew how they were defined and measured, too. For instance,
your drawing certainly has 4.5 pF CM capacitance, but if you really
drove it differentially, e.g. with a transformer, it'd have 1.825 pF
differential capacitance. With a pi network there's no way for the
differential Cin to be less than a quarter of the CM Cin.
I thought about the following possibilities:
1. C_gs of the two input devices appearing in series for differential,
but not for CM. Whoops, that should be worse for differential--it's
bootstrapped for CM.
2. Capacitance of current source driving the sources. Right sign this
time, but if (1) is small, this has to be small too because it's in
series with C_gs.
3. Miller capacitance caused by non-ideal behaviour of cascodes. Also
wrong sign.
But the possibility that strikes me is
4. They've cross-coupled the gate of each input device to the opposite
device's drain to get some negative Miller capacitance, sort of like
unilateralizing a push-pull RF power amp. From a circuits point of
view that would look like a negative differential capacitance, which
might explain how C_diff can be less than C_cm/4. It doesn't help me
any in building low-light TIAs, because it's the single-ended
capacitance that matters for loop stability with high R_F and so on. I
don't see how it helps anything else either, since even when building
noninverting amps feedback forces the input swings to be nearly all CM.
Mysterious.
The reason this is rotting my socks just now is that National
discontinued the LF357 in 2004 (*requiescat in pace*), and there's
nothing available with its combination of FET input, +-15V supplies, 2
pF C_in, 12 nV noise, and 15 MHz GBW. The OPA656 is a drop-in
replacement but only runs on +-5, so you lose 10 dB dynamic range (it
also costs $5 vs $0.75). The OPA627 has low noise and enough speed, but
it also has 7.5 pF input capacitance (8 differential, 7 CM). That
forces me to drop the feedback resistor by a factor of almost 3, and
loses me a decibel of SNR, while costing $18.
This is _after_ the photodiode capacitance has been fixed by the
bootstrapped cascode, mind---all I want is for the stupid amplifier to
get out of its own way. Grrr.
As Mehitabel would put it, "Toujours gai, toujours gai."
Cheers,
Phil Hobbs
gghe...@gmail.com
I saw a simple circuit from Bob Pease for "rolling your own" JFET
front end for TIA's. It's printed out and tucked into a folder
somewhere.
George
Phil Hobbs
>
> Phil Hobbs wrote:
>> The reason this is rotting my socks just now is that National
>> discontinued the LF357 in 2004 (*requiescat in pace*), and there's
>> nothing available with its combination of FET input, +-15V supplies, 2
>> pF C_in, 12 nV noise, and 15 MHz GBW. The OPA656 is a drop-in
>> replacement but only runs on +-5, so you lose 10 dB dynamic range (it
>> also costs $5 vs $0.75). The OPA627 has low noise and enough speed, but
>> it also has 7.5 pF input capacitance (8 differential, 7 CM). That
>> forces me to drop the feedback resistor by a factor of almost 3, and
>> loses me a decibel of SNR, while costing $18.
>>
>> This is _after_ the photodiode capacitance has been fixed by the
>> bootstrapped cascode, mind---all I want is for the stupid amplifier to
>> get out of its own way. Grrr.
>>
>> As Mehitabel would put it, "Toujours gai, toujours gai."
>>
>> Cheers,
>>
>> Phil Hobbs
>
> I saw a simple circuit from Bob Pease for "rolling your own" JFET
> front end for TIA's. It's printed out and tucked into a folder
> somewhere.
>
Thanks. A couple of BF862s will do a good job at AC, true. The problem
is the amount of extra crapola you have to hang on them to get decent DC
behaviour. (Also one begins to wonder why one bothers with an op amp at
that point.)
The basic issue is that 300K resistors are so very noisy compared to
almost anything else, so you have to use really big ones to overcome
their Johnson noise, which in turn leads to all sorts of bandwidth
problems like this.
I have a new design idea based on using all current feedback that more
or less avoids this problem by making the feedback currents very quiet.
(This is done by dropping lots of voltage across the resistors in the
current sources.) Still a partly-baked idea at this point, but it
should be possible to do a better job with a simpler circuit that way.
Cheers,
Phil Hobbs
John Larkin
<pcdhSpamM...@electrooptical.net> wrote:
It is if you do a 3-terminal C measurement. My old green Boonton
c-meter would measure that 0.7 pF accurately and ignore much bigger
caps to ground. So it depends on the capacitance model they used but
didn't define.
John
Phil Hobbs
Interesting. I've never played with one, but given the very strong
tendency to believe test equipment, that might well be it. I suppose
the only thing to do is get out the old soldering iron and see--and here
I was trying to mail the MS tomorrow.
So I suppose one good way to measure C_CM would be to build a zero-gain
amplifier (a diff amp with the two inputs tied together), but with much
higher impedance in the + arm. Feedback would pretty well bootstrap out
C_diff, so I should just get sin(2 pi f R C_CM).
Using the loop behaviour to estimate C_diff would require quite a bit of
curve fitting. Any suggestions on how to measure that?
Cheers,
Phil
John Larkin
<pcdhSpamM...@electrooptical.net> wrote:
[Oh, yeah, Rob looked at the thing - he's smarter than I am - and it
looks fine. I'd like you to discuss *fast* (GHz maybe) photodiode amps
at some point... 3e maybe. I'm certain we're doing it wrong.]
>
>So I suppose one good way to measure C_CM would be to build a zero-gain
>amplifier (a diff amp with the two inputs tied together), but with much
>higher impedance in the + arm. Feedback would pretty well bootstrap out
>C_diff, so I should just get sin(2 pi f R C_CM).
>
>Using the loop behaviour to estimate C_diff would require quite a bit of
>curve fitting. Any suggestions on how to measure that?
>
Open-loop, it's easy. Poke an AC voltage into one input, and mesaure
the current that flows to ground from the other. That's what the
Boonton does. It outputs (I recall) 20 millivolts at 1 MHz on the
driven side, and has a low-impedance, high gain tuned amp and a
synchronous detector on the sense side.
http://www.slack.com/images/TE/Boonton72B.jpg
http://www.used-line.com/cgi-bin/a_view/c_pic.cfm?ItemId=6276877
The low range is 1 pF full scale, and it has a bias input on the back,
so it's cool for characterizing diodes and such.
John
Phil Hobbs
> On Tue, 30 Sep 2008 23:07:19 -0400, Phil Hobbs
> <pcdhSpamM...@electrooptical.net> wrote:
>>> It is if you do a 3-terminal C measurement. My old green Boonton
>>> c-meter would measure that 0.7 pF accurately and ignore much bigger
>>> caps to ground. So it depends on the capacitance model they used but
>>> didn't define.
>>>
>> Interesting. I've never played with one, but given the very strong
>> tendency to believe test equipment, that might well be it. I suppose
>> the only thing to do is get out the old soldering iron and see--and here
>> I was trying to mail the MS tomorrow.
Thanks. (Thanks to you too, Rob.) I'll try fleshing the partly-baked
idea out a bit and post something if it's interesting looking. Might be
worth a short article if so--it looks a dB or two better than the
bootstrapped cascode even, and it might help get round the op amp drought.
>
> [Oh, yeah, Rob looked at the thing - he's smarter than I am - and it
> looks fine. I'd like you to discuss *fast* (GHz maybe) photodiode amps
> at some point... 3e maybe. I'm certain we're doing it wrong.]
That would be a bit of a learning experience for me too--I've been a bit
spooked by the guys I know who design TIA chips and so on...very fast,
all 50 ohms and 20 pA 1-Hz current noise, with PDs integrated on the
chip for the most part. However, there's probably an interesting niche
in the 1-2 GHz range that they don't care about anymore, so I can build
little prototypes without running into any more scorn and derision than
usual. There's more high frequency stuff in this edition, but it's
mostly reactive matching networks, Bode's bandwidth vs return loss
theorem, and so on.
I'd love to know how Miteq makes those amps with the 25 K noise
temperature. They work at 77 K, so they have to be FETs of some
sort---they aren't parametric amps, for sure. If I can find an excuse
to buy one, I might buy two and can-open the spare.
>> So I suppose one good way to measure C_CM would be to build a zero-gain
>> amplifier (a diff amp with the two inputs tied together), but with much
>> higher impedance in the + arm. Feedback would pretty well bootstrap out
>> C_diff, so I should just get sin(2 pi f R C_CM).
>>
>> Using the loop behaviour to estimate C_diff would require quite a bit of
>> curve fitting. Any suggestions on how to measure that?
>>
>
> Open-loop, it's easy. Poke an AC voltage into one input, and mesaure
> the current that flows to ground from the other. That's what the
> Boonton does. It outputs (I recall) 20 millivolts at 1 MHz on the
> driven side, and has a low-impedance, high gain tuned amp and a
> synchronous detector on the sense side.
>
Yeah, I was afraid of that. I sort of doubt the capacitances will be
the same with the amp unpowered or railed, don't you?
> http://www.slack.com/images/TE/Boonton72B.jpg
Cute. Looks useful, too.
>
> http://www.used-line.com/cgi-bin/a_view/c_pic.cfm?ItemId=6276877
>
>
> The low range is 1 pF full scale, and it has a bias input on the back,
> so it's cool for characterizing diodes and such.
>
Cheers,
Phil Hobbs
John Larkin
<pcdhSpamM...@electrooptical.net> wrote:
My problem with the commercial TIA chips is that their gain is so high
that they saturate at sub-mw levels, and worse that they all have AGC.
AGC is hell on calibrated, dc-coupled receivers. All telecom stuff,
basically. We just did a pretty clean 1 GHz o/e converter, made from
regular parts on boards, 2 mw full range, but the noise level is going
to be horrible compared to the telecom pin-tia gadgets.
>
>I'd love to know how Miteq makes those amps with the 25 K noise
>temperature. They work at 77 K, so they have to be FETs of some
>sort---they aren't parametric amps, for sure. If I can find an excuse
>to buy one, I might buy two and can-open the spare.
Fet distributed amps? Nf improved by the square root of some silly
value of N?
>
>>> So I suppose one good way to measure C_CM would be to build a zero-gain
>>> amplifier (a diff amp with the two inputs tied together), but with much
>>> higher impedance in the + arm. Feedback would pretty well bootstrap out
>>> C_diff, so I should just get sin(2 pi f R C_CM).
>>>
>>> Using the loop behaviour to estimate C_diff would require quite a bit of
>>> curve fitting. Any suggestions on how to measure that?
>>>
>>
>> Open-loop, it's easy. Poke an AC voltage into one input, and mesaure
>> the current that flows to ground from the other. That's what the
>> Boonton does. It outputs (I recall) 20 millivolts at 1 MHz on the
>> driven side, and has a low-impedance, high gain tuned amp and a
>> synchronous detector on the sense side.
>>
>Yeah, I was afraid of that. I sort of doubt the capacitances will be
>the same with the amp unpowered or railed, don't you?
Build a regular Ri/Rf inverting amp. Crank the input level and
frequency up until you have significant swing at the - input. Measure
the ac current that sloshes out of the grounded + input, with another
opamp maybe, or maybe just a spectrum analyzer. Just make sure the
rails are very stiff.
John
gghe...@gmail.com
<pcdhSpamMeSensel...@electrooptical.net> wrote:
>
> - Show quoted text -
"Thanks. A couple of BF862s will do a good job at AC, true. The
problem
is the amount of extra crapola you have to hang on them to get decent
DC
behaviour. (Also one begins to wonder why one bothers with an op amp
at
that point.)"
Yup, it would be getting very busy.
"The basic issue is that 300K resistors are so very noisy compared to
almost anything else, so you have to use really big ones to overcome
their Johnson noise, which in turn leads to all sorts of bandwidth
problems like this."
Hmm, could you use some sort of coorelation scheme to reduce the
noise? If you had two TIA's running off the same PD current you could
generate two signals Vs+Vn1 and Vs+Vn2, where Vs is the common signal
and the Vn's are the uncorrelated noises from each amp. Now if I had
a "common mode" amplifier, (as opposed to a differential amp), I could
amplify the "common" signal and not the noise... Unfortunately the
only thing I can think of is summing the two signals. Which "I
think" gives me a 1.414 increase in the SNR. Is there any way to do
better?
" I have a new design idea based on using all current feedback that
more
or less avoids this problem by making the feedback currents very
quiet.
(This is done by dropping lots of voltage across the resistors in
the
current sources.) Still a partly-baked idea at this point, but it
should be possible to do a better job with a simpler circuit that
way."
Ahh do tell, or post a message/link if things pan out. Are you
talking about using CFA's or some other configuration? I've always
wanted an excuse to learn about CFA's.
George Herold
Phil Hobbs
This works for voltage noise, as in the two-point correlation method,
which does something similar--you measure the voltage twice and
cross-correlate the two measurements. It relies on time-averaging, so
you need stationary statistics. Thus you can do a great job measuring
noise, but not so great measuring signal.
An ensemble average, where you use 10 voltmeters, form the 45 unique
pairwise products, and average those, relaxes the requirement for
stationary statistics, but still measures the squared magnitude.
You win linearly with the number of voltmeters, because the improvement
is sqrt(#pairs) and the number of pairs goes as N**2. Note that it only
works for voltage noise--current noise is a real current that comes out
the input terminals of each amplifier, so they all measure the sum of
their noise currents, and this will survive the averaging since it's
indistinguishable from signal.
Making a linear measurement isn't possible that way, unfortunately. You
can get 3 dB reduction in the circuit noise by connecting a TIA to each
end of the PD, as you suggest (which is an interesting idea). You'd
actually want to subtract the results, because the two currents are
equal and opposite. It requires some fancy footwork on the power
supplies, of course, because you lose much more than 3 dB by not
reverse-biasing the diode.
> " I have a new design idea based on using all current feedback that
> more
> or less avoids this problem by making the feedback currents very
> quiet.
> (This is done by dropping lots of voltage across the resistors in
> the
> current sources.) Still a partly-baked idea at this point, but it
> should be possible to do a better job with a simpler circuit that
> way."
>
> Ahh do tell, or post a message/link if things pan out. Are you
> talking about using CFA's or some other configuration? I've always
> wanted an excuse to learn about CFA's.
> Nothing magical. It starts from the observation that BJTs are very
quiet from a voltage gain perspective (T_noise ~ Tj/beta) but look like
150 kelvin resistors from a current gain perspective, like all ideal
diodes. So if you pick a superbeta transistor and connect the PD to its
base (like a phototransistor), you're stuck with 150 kelvin T_N and
significant nonlinearity. On the other hand, if you enclose it in a
current-feedback loop that pulls all but epsilon times the photocurrent
out of the base circuit, and optimize epsilon, you can win overall noise
temperature like sqrt(beta). All the feedback currents have to be way
sub-Poissonian, so you have to drop a lot of voltage across the emitter
resistors of the current sources, but in principle with a beta of 1000,
you can make a TIA whose noise temperature is of the order of 10 kelvin,
with a reasonably favourable tradeoff of noise vs bandwidth.
Vanilla CFB amps were initially puzzling to me back in the day, because
their bandwidth is not a strong function of the feedback gain. It
became a lot clearer when I imagined making a summing amp from a regular
VFB op amp--if you put a small resistor between the summing junction and
ground, the VFB amp's bandwidth becomes gain-independent as well. (CFBs
do this without the horrible noise degradation that the stupid resistor
would cause.)
Cheers,
Phil Hobbs
gghe...@gmail.com
wrote:
>
> - Show quoted text -
"Making a linear measurement isn't possible that way, unfortunately.
You
can get 3 dB reduction in the circuit noise by connecting a TIA to
each
end of the PD, as you suggest (which is an interesting idea). You'd
actually want to subtract the results, because the two currents are
equal and opposite. It requires some fancy footwork on the power
supplies, of course, because you lose much more than 3 dB by not
reverse-biasing the diode. "
Yes I was thinking about this last night. As you say you need to
hook a TIA to each end of the PD, and then you will have to subtract
(and not add) the two signals. And it looks like you lose the
ability to bias the PD. But I "think" I've seen the circuit trick to
beat that. (Though I haven't tried it.) So the non-invering input of
one TIA is grounded, and then you make the non-inverting input of the
other TIA the bias voltage. This does limit your bias voltage to be
less than the op-amp supply voltage, but I don't think that is much of
a constraint. It does look like it might limit the dynamic range a
bit....I'd have to draw all this out. It seems like one could "play
games" with where the ground was defined and still retain most of
dynamic range.
And Thanks for the CFB lesson... I'll think about it.
George Herold
Phil Hobbs
It's potentially a nice solution to the +-5V restriction on many of the
good FET op amps e.g. the OPA656.(*) You can run the op amps off +15/+5
and -5/-15 (suitably protected against power supply sequencing funnies,
loss of ground, and so on). You can do the level shifting like this:
----------*--RFRFRF-------*---R1R1R1--- +15
| | |
| | +15 |
| | |\ | |
| | | \| |
| ---|- \ | /
| | \ |<
| | >-----|
| | / |\
| +10---|+ / | \
| | /| | | -(1+R_F/R_1)*I_photo +I_bias
| |/ +5 | V
| |
| *------
| | |
| R |
| R |
----- R ------
/ \ R | - |
/___\ | | |
| GND | |----0 Out (plus bias)
| | | |
| R | + |
| R ------
| R |
| R |
| | |
| -5 *------
| |\ | |
| | \| | | (1+R_F/R_1)*I_photo - I_bias
| -10---|+ \ | / V
| | \ |/
| | >-----|
| | / |>
| ---|- / | \
| | | /| |
| | |/ -15 |
| | |
----------*--RFRFRF-------*---R1R1R1--- -15
This gets you back some of the dynamic range you lose by using the +-5V
op amps. Also, the current multiplication relaxes the noise
requirements on the subsequent stages.
Cheers,
Phil Hobbs
Put BJT follower in the feedback path (base to op amp output, emitter
driving R_F), which produces (for level shifting by driving the base
with the op ampwith a BJT in series with each feedback resistor
(*) TI wants everybody to use the 657, correctly pointing out that
almost all TIAs run at some gigantic noise gain on account of the PD
capacitance, so that you don't have to worry about using a decompensated
op amp even though it looks like a unity gain application. The thing
they don't tell you is that the 657 has twice the input capacitance of
the 656, so it's worse at low photocurrents.
John Larkin
On our list of things to do is a really fast DC-coupled o/e converter.
One end of the photodiode goes into a blinding-fast AC-coupled amp,
and the other end to a slow DC amp. Combine them later.
The real bear is to find a good way to calibrate the fast gain path.
Trimpots have been demonstrated to not work!
John
Jim Thompson
On Fri, 03 Oct 2008 09:58:50 -0700, John Larkin
<jjla...@highNOTlandTHIStechnologyPART.com> wrote:
[snip]
>
>
>On our list of things to do is a really fast DC-coupled o/e converter.
>One end of the photodiode goes into a blinding-fast AC-coupled amp,
I presume there's a DC path at the input to the "blinding-fast
AC-coupled amp" ??
>and the other end to a slow DC amp. Combine them later.
>
>The real bear is to find a good way to calibrate the fast gain path.
>Trimpots have been demonstrated to not work!
>
>John
>
>
...Jim Thompson
John Larkin
<To-Email-Use-Th...@My-Web-Site.com> wrote:
>
>On Fri, 03 Oct 2008 09:58:50 -0700, John Larkin
><jjla...@highNOTlandTHIStechnologyPART.com> wrote:
>
>[snip]
>>
>>
>>On our list of things to do is a really fast DC-coupled o/e converter.
>>One end of the photodiode goes into a blinding-fast AC-coupled amp,
>
>I presume there's a DC path at the input to the "blinding-fast
>AC-coupled amp" ??
Do you mean some way to keep the electrons from piling up on one side
of the pd? Sure.
Or do you mean that the initial coupling, into the first fast gain
stage, should be DC coupled? That would be fine... the photodiode
current will be low, so a mmic or whatever wouldn't mind the dc
component.
MMICs have horrendous input and output dc offsets, more horrendous
over temperature, so it makes sense to ac couple the fast path.
We actually used opamps on the one we just finished, all DC coupled at
1 GHz net o/e bandwidth. But that's about the end of the line for
opamps.
John
Phil Hobbs
> On our list of things to do is a really fast DC-coupled o/e converter.
> One end of the photodiode goes into a blinding-fast AC-coupled amp,
> and the other end to a slow DC amp. Combine them later.
>
> The real bear is to find a good way to calibrate the fast gain path.
> Trimpots have been demonstrated to not work!
>
Why not let the fast path do what it likes, and dink the slow path to
match?
If you have a shot-noise calibration, you can just announce what the
scale factor is. Systems like that always have to have overall
calibrations anyway.
Cheers,
Phil Hobbs
Joerg
[...]
> On our list of things to do is a really fast DC-coupled o/e converter.
> One end of the photodiode goes into a blinding-fast AC-coupled amp,
> and the other end to a slow DC amp. Combine them later.
>
> The real bear is to find a good way to calibrate the fast gain path.
> Trimpots have been demonstrated to not work!
>
Servoed PIN diodes? Of course then you'll have to find nice dual PIN
diodes with the desired minimum carrier lifetime, low enough capacitance
and such. If you find suitable ones you could do this:
One goes into the RF path. The other goes into a separate amp and a
pilot signal of precise amplitude is injected. This feeds some chip that
has a good RSSI output and there is your servo signal you can use to
offset for drift in the PIN diodes.
Depending on how drifty your stuff is and thus how much calibration
range is required you may need more than one pair.
--
Regards, Joerg
http://www.analogconsultants.com/
"gmail" domain blocked because of excessive spam.
Use another domain or send PM.
Joerg
> John Larkin wrote:
>
>> On our list of things to do is a really fast DC-coupled o/e converter.
>> One end of the photodiode goes into a blinding-fast AC-coupled amp,
>> and the other end to a slow DC amp. Combine them later.
>>
>> The real bear is to find a good way to calibrate the fast gain path.
>> Trimpots have been demonstrated to not work!
>>
>
> Why not let the fast path do what it likes, and dink the slow path to
> match?
>
But then the whole overall gain stability would go to pots. In my last
app any more than 0.5dB would have caused them to be displeased with
their electronics guy (me...).
> If you have a shot-noise calibration, you can just announce what the
> scale factor is. Systems like that always have to have overall
> calibrations anyway.
>
Which makes me wonder. Aren't there any photodiodes that have a built-in
LED with defined output intensity so you could calibrate them? Maybe in
a quiet phase, or continuous duty if you could modulate it in a band you
aren't using. Kind of like laser diodes with back facet photodiodes,
just the other way around.
John Larkin
<pcdhSpamM...@electrooptical.net> wrote:
>John Larkin wrote:
>
>> On our list of things to do is a really fast DC-coupled o/e converter.
>> One end of the photodiode goes into a blinding-fast AC-coupled amp,
>> and the other end to a slow DC amp. Combine them later.
>>
>> The real bear is to find a good way to calibrate the fast gain path.
>> Trimpots have been demonstrated to not work!
>>
>
>Why not let the fast path do what it likes, and dink the slow path to
>match?
Because I'd like to deliver 1 volt per milliwatt, calibrated.
We can do it by soldering selected resistors, but that's inelegant.
John
Joerg
Resistors plus one or two of these maybe?
http://semicon.njr.co.jp/njr/hp/fileDownloadMedia.do?_mediaId=797
gghe...@gmail.com
Ahh, This is more clever than what I had in mind. Say is there some
trick to viewing the ASSC (sp) art? It seems like all the important
spaces are left out of my view.
George Heorld
Phil Hobbs
>> almost all TIAs run at some gigantic noise gain on account of the PD
>> capacitance, so that you don't have to worry about using a decompensated
>> op amp even though it looks like a unity gain application. The thing
>> they don't tell you is that the 657 has twice the input capacitance of
>> the 656, so it's worse at low photocurrents.
>
>
> Ahh, This is more clever than what I had in mind. Say is there some
> trick to viewing the ASSC (sp) art? It seems like all the important
> spaces are left out of my view.
>
> George Heorld
Switch to a monospaced font, e.g. Courier.
Cheers,
Phil Hobbs
???
the circuit?
"Phil Hobbs" <pcdhSpamM...@electrooptical.net>
??????:48E25C23...@electrooptical.net...