Ferrite Core Design

[Bradley Zweig]

ThePower Chart characterizes the power handling capacity of each ferrite core based upon the frequency of operation, the circuit topology, the flux level selected, and the amount of power required by the circuit. If these four specifics are known, the core can be selected from the Typical Power Handling Chart.

The power handling capacity of a transformer core can also be determined by its WaAc product, where Wa is the available core window area, and Ac is the effective core cross-sectional area. Using the equation shown below, calculate the WaAc product and then use the Area Product Distribution (WaAc) Chart to select the appropriate core.

The MDT software allows application-related parameters to be calculated for all available EPCOS ferrite cores and / or materials. It provides access to their digitized material data including their graphical representations. The user manual of the software contains a detailed description of all functions. The tool can be used as online version, tested in browsers like Google Chrome, Mozilla Firefox, Internet Explorer and Opera, or as download, developed for Windows 7 and 10. The online version does not offer the same comprehensive set of functions as the desktop version.

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The following design guide may also be downloaded as a PDF. For other inquiries regarding inductor design with Magnetics ferrite cores, Contact our Applications Engineers or submit a Custom Inductor Design request.

Ferrite E cores and pot cores offer the advantages of decreased cost and low core losses at high frequencies. For switching regulators, power materials are recommended because of their temperature and DC bias characteristics. By adding air gaps to these ferrite shapes, the cores can be used efficiently while avoiding saturation.

These core selection procedures simplify the design of inductors for switching regulator applications. One can determine the smallest core size, assuming a winding factor of 50% and wire current carrying capacity of 500 circular mils per ampere.

2. Locate the LI value on the Ferrite Core Selector chart below.

Follow this coordinate in the intersection with the first core size curve. Read the maximum nominal inductance, AL, on the Y-axis. This represents the smallest core size and maximum AL at which saturation will be avoided.

The above curves represent the locus of points up to which effective permeability remains constant. They show the maximum allowable DC bias, in ampere-turns, without a reduction in inductance. Beyond this level, inductance drops rapidly.

Before proceeding, the implications of the decision to use ferrite cores rather than any other material needs to be considered. Make sure this is the best material for your application before proceeding.

The design we are using as the basis of this article is aiming for an inductance of about 1 mH, and the ferrite material that will be considered is a planar type made by Ferroxcube. The part number is E38/8/25, and two halves are needed to make a full core-set:

On the right is how the core set and PCB will be constructed; The copper PCB tracks will form the coil loops, and we may have to stack several small PCBs to get the right number of turns. Another aim of this design is to keep the overall profile height as low as possible. Hence, we have targeted a planar core-set.

The E38/8/25 core can be bought in several different ferrite materials. The common material types are 3C90 and 3F3. The next thing is to look into both these material types to see if one might be preferred over the other. The first comparison is the frequency response, i.e., how high a frequency is the ferrite material suitable for:

Because ferrite materials concentrate the magnetic flux and (pretty much) ensure that all winding turns are coupled to each other, the relationship between the number of turns and inductance is, therefore:

Of course, if you have a different application circuit in mind for the wound component, then the current calculation may be more straightforward than that shown above. In either case, you still need to calculate the peak current to see if there might be a potential core saturation problem.

We calculated 0.918 amps for our target application, and we know that this current is the transformer ferrite core magnetization current for a flyback converter. Therefore, it could easily over-saturate the core.

This is the predicted peak flux density, and we know from the literature that it is too much for ferrite cores. Ferrite will heavily saturate around 0.4 teslas, so we have to slightly rethink what we are doing.

The simple answer is yes, we need to reduce the predicted peak flux density. The good news is that we can buy the E38/8/25 core set already pre-gapped. Do you recall this data sheet extract from stage 4?

The numbers tally reasonably well. The gapped core-set with an H field of 566.1 has a simulated flux density of 129 mT, whereas the hand-calculation arrived at 153 mT. For the ungapped 12-turn core-set with an H field of 210.2, we get a simulated flux density of 413 mT (very close to the hand calculation of 414.8 mT).

Mark Harris is an engineer's engineer, with over 16 years of diverse experience within the electronics industry, varying from aerospace and defense contracts to small product startups, hobbies and everything in between. Before moving to the United Kingdom, Mark was employed by one of the largest research organizations in Canada; every day brought a different project or challenge involving electronics, mechanics, and software. He also publishes the most extensive open source database library of components for Altium Designer called the Celestial Database Library. Mark has an affinity for open-source hardware and software and the innovative problem-solving required for the day-to-day challenges such projects offer. Electronics are passion; watching a product go from an idea to reality and start interacting with the world is a never-ending source of enjoyment.

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For my design, I have used Bmax as 35mT, or 3500 gausses. I have calculated primary using this formula.(0.5 * Vin * 10^8)/ 4 * F * Bmax * Cross-section-Area). I have added 0.5 cause my design is Half Bridge. My switching Frequency is 50Khz. And my cross-section area is 2.9cm^2. So the result is 3.69 turn, so have used 4 turns in primary using 6mm square Wire. I need not more than 28mA in Secondary, so I have used 37AWG wire. And the secondary turn I have calculated using *(Vs/Vp)Np= 2666 Turns in secondary. But I have used 3000 turns in secondary.

Now the problem I am facing is while my secondary is at 0/NO loads,my Primary current is drawing 15-ampere peak at 40 volts RMS. When I am trying to give more voltage in the primary, the primary is drawing more current proportional to the primary voltage. Even when I am trying to generate an ARC in secondary the primary peak current is the remain the same. At 40 volt RMS my secondary should generate 4*666.66 = 26640 Volt. But the arc is only 1cm Long. My power is supply can provide up to 3KW.

Why my ferrite transformer drawing 15amp peak at 40-volt RMS at NO loads ?, I mean when the secondary is not generating any ARC or the secondary wires are not even close? I have measured the secondary wire resistance. The resistance is 535 ohms.

If you are trying to get a 100 kV DC output then limit your transformer to producing an RMS output in the mid-kV range i.e. 3 to 5 kV AV and then use a Cockcroft-Walton voltage multiplier on the output that is oil-immersed and I don't mean cooking oil.

The reason I point out about limiting the transformer AC output is that with the number of turns needed, the insulation between secondary layer and the leakage inductance, you'll just about avoid hitting the self-resonant frequency of the transformer. If you hit SRF then you'll get really big problems that you'll never control.

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